Method and apparatus for an active negative-capacitor circuit to cancel the input capacitance of comparators

ABSTRACT

The differential output of a Programmable Gain Amplifier (PGA) is loaded by the input differential gate capacitance of a plurality of Analog to Digital convertors (ADC) comparators and the differential metal layer traces to interconnect these comparators to the PGA. The differential capacitive load presented to the PGA is quite large and reduces the bandwidth of this interconnect between the PGA and ADC. To overcome the performance degradation due to the differential capacitive load, an active negative-capacitor circuit cancels the effect of the large input capacitance of the ADC comparators. This cancelation extends the gain characteristics of the interconnect between the PGA&#39;s output and the inputs of the first stage of the comparators. The active negative-capacitance is comprised of a cross pair NMOS with a capacitor connecting their sources where each NMOS is biased by a current source.

CROSS-REFERENCE TO RELATED APPLICATIONS

This continuation application claims the benefit of the priornonprovisional U.S. application Ser. No. 13/602,216, filed Sep. 3, 2012,entitled “Method and Apparatus for an Active Negative-Capacitor Circuitto Cancel the Input Capacitance of Comparators” invented by the sameauthor as the present application and is related to the U.S. applicationentitled “Method and Apparatus for Reducing the Clock Kick-Back of ADCComparators While Maintaining Transistor Matching Behavior” filed onSep. 3, 2012, which is invented by the same inventor as the presentapplication, both applications are assigned to the same assignee andincorporated herein by reference in their entireties.

BACKGROUND OF THE INVENTION

The Federal Communications Commission (FCC) has allotted a spectrum ofbandwidth in the 60 GHz frequency range (57 to 64 GHz). The WirelessGigabit Alliance (WiGig) is targeting the standardization of thisfrequency band which will support data transmission rates up to 7 Gbps.Integrated circuits, formed in semiconductor die, offer high frequencyoperation in this millimeter wavelength range of frequencies. Some ofthese integrated circuits utilize Complementary Metal OxideSemiconductor (CMOS), Silicon-Germanium (SiGe) or Gallium Arsenide(GaAs) technology to form the dice in these designs. Since WiGigtransceivers use Digital to Analog Converters (DAC), the reduced powersupply impacts the performance of the DAC's.

Complementary Metal Oxide Semiconductor (CMOS) is the primary technologyused to construct integrated circuits. N-channel transistors andP-channel transistors (MOS transistor) are used in this technology whichuses fine line technology to consistently reduce the channel length ofthe MOS transistors. Some of the current values for this technologyinclude the channel length being 40 nm, the power supply of VDD equaling1.2V and the number of layers of metal levels being 8 or more. Thistechnology typically scales with technology.

CMOS technology delivers a designer the ability to form a very largesystem level design on one die which is known as a System On a Chip(SOC). The SOC is a complex system with millions, if not billions, oftransistors which contain analog circuits and digital circuits. Theanalog circuits operate purely analog, the digital circuits operatepurely digital and these two circuits types can be combined together toform circuits operating in a mixed-signal mode.

For example, digital circuits in their basic form only use digital logicand some examples can be a component comprising at least one; processor,memory, control logic, digital I/O circuit, reconfigurable logic and/orhardware programmed that to operate as hardware emulator. Analogcircuits in their basic form use only analog circuits and some examplescan be a component comprising at least one; amplifier, oscillator,mixer, and/or filter. Mixed signal in their basic form only use bothdigital and analog circuits and some examples can be a componentcomprising at least one: Digital to Analog Converter (DAC), Analog toDigital Converter (ADC), Programmable Gain Amplifier (PGA), Power Supplycontrol, Phase Lock Loop (PILL), and/or transistor behavior control overProcess, Voltage and Temperature (PVT). The combination of digital logiccomponents with analog circuit components can appear to behave likemixed signal circuits; furthermore, the examples that have been providedare not exhaustive as one knowledgeable in the arts understands.

One of the critical design parameters of a transceiver occurs when acontinuous analog signal is converted into a digital time signal in theADC. A flash ADC uses a linear reference voltage source that is tappedand applied to one of the differential inputs of a number of parallelcomparators. The input analog value is applied to the other differentialinput of all of the comparators simultaneously providing a very quickcomparison. Several critical issues can occur in this conversion whichincludes: 1) the matching of the input transistors within and betweenthe comparators; 2) clock kick-back from the clock enabling thecomparators to the input signal; and 3) a reduction in bandwidth betweenthe PGA and the large capacitive load of the ADC and the interconnect.

The matching of transistors within and between the comparators usesdummy transistors which use up valuable semiconductor area and causes anincrease in the power dissipation due to increased wire lengths of thedata and clock lines. If the matching of the transistors is notmaintained well, the issue becomes a mismatching condition. Thebandwidth of the ADC is limited by the input signal driving the inputcapacitive load of all the parallel comparators and the interconnect.This necessitates that the transistor width of the input transistors ofthe comparators to have an upper bound. Such a transistor width may notbe sufficient and cause the matching problem to become more severe.Increasing the width of the transistor beyond this upper bound helpsovercome the mismatching condition but causes the bandwidth of the ADCto reduce. Other solutions are required to resolve the mismatchingcondition yet allow the desired bandwidth to be satisfiedsimultaneously.

Clock kick-back from the clock to the input signal of a gate usuallyoccurs via the capacitance coupling between the terminals of the activetransistor, i.e., the gate overlap capacitance from the source and drainterminals to the gate terminal of an MOS transistor. As the width of thetransistor is increased, the coupling capacitance increases whichincreases the clock kick-back. In addition, the power dissipation of thesystem increases as well because of the increased width of thetransistors. A second aspect of clock kick-back is the transientbehavior of the circuit being clocked between an initialization stateand a steady state. The internal nodes of the clocked circuit during thetransient period also generate a clock kick-back besides increasing thedelay of the operation of the circuit. Several solutions are providedwhich overcome these shortcomings by reducing clock kick-back, therebyimproving the performance of the circuit.

The signal delivery between the PGA and the ADC can be delayed by thelarge gate capacitance of the MOS transistors and the interconnectcapacitance of the metal trace used to couple these comparators. Thetransfer of signals between the PGA and the ADC causes a decrease in thebandwidth of the path due to the capacitance. Typically, the performanceof the ADC can be improved by increasing the width of the transistors toachieve a faster response. But the larger transistors, besidesincreasing the kick-back and power dissipation, also increase the delayof the signal delivery because of the larger gate capacitance beingpresented to the output of the PGA. The transfer of data on theinterconnect between the PGA and ADC is critical to improving theperformance of the system. A new technique will be presented to improvethe signal's transfer at this critical node and improve the bandwidth ofthe captured signal.

BRIEF SUMMARY OF THE INVENTION

As the power supply voltage continues to reduce in the scaling ofintegrated circuits, the voltage headroom for analog integrated circuitsdecreases correspondingly. This makes the design of high performancecircuits such as ADC systems in an integrated circuit much morechallenging. Thus, the proper layout of comparators in an integratedcircuit implementing an ADC is of great importance to overcome themismatching condition. The comparators require that criticaltransistors, such as the input transistors, are matched to one anotherin terms of their local environment. The layout features of adjacentcircuits can impact the behavior of a transistor in the current circuitby forming mismatches due to the various processing steps used tomanufacture the integrated circuit. Increasing the length of the inputtransistors of the comparators helps to overcome this mismatchcondition, but the input capacitance of the comparators increases andthe performance decreases causing the bandwidth of the ADC to bereduced.

Mismatches between transistors, especially input transistor pairs, willlead to false comparator outputs. In addition, improper layout may leadto significant mismatches both within one comparator and betweenidentical comparators during manufacture. Both mismatches may result inthe ADC making false decisions. Furthermore, a poorly packed layoutdesign will add unnecessary interconnect trace lengths for both signaland clock, especially for an ADC with large number of comparators. Theselonger trace interconnects, implemented as a differential signal thathas parallel routings, will dramatically degrade the bandwidth of thesystem. The longer clock routing requires larger clock buffers whichincrease the overall power consumption, and more profoundly, additionalclock jitter will occur, which will cause problems such as bubbles inthe decoded results and lower ENOB (Effective Number of Bits). Thelonger power supply lines introduce additional IR drops, which furthercontribute to the mismatches among different comparators.

A simple layout technique is proposed in this embodiment that minimizesthe concerns of the above aspects. Instead of having extra dummy fingersfor each comparator to reduce the mismatch within one comparator betweeninput transistors, the comparators are aligned right next to each other,so that all the input transistors share one whole active area. Thereby,the fingers on the edge of the active area serve as the dummy fingersfor the neighboring comparators.

In another embodiment, the core concept of this ADC is the high-speedfully-differential comparators which are clocked at 2640 MHz that isused in the 60 GHz transceiver. Basically, each comparator consists offour parts: a pre-amplifier stage which samples and amplifies the inputsignal from the preceding stage or Programmable Gain Amplifier (PGA); acapture stage that is clocked to capture the contents of thepre-amplifier stage; a pair of clocked cross-coupled transistors whichregenerates the small signal to nearly a rail-to-rail signal; and anoutput latch which latches up the previous results after regenerationfor application to the following static CMOS circuits. The pre-amplifierstage is not clocked; therefore, the pre-amplifier stage does not sufferinitialization and transient behavior effects. Instead, a capture stageuses the clock signal to transfer the contents of the pre-amplifierstage into a memory regeneration stage. The capture stage is clocked bypulses that are timed to minimize the clock kick-back generated by thememory regeneration stage.

In another embodiment, since a flash ADC converter is used, a number ofcomparators are placed in parallel with their inputs connected together.The clock kick-back from the many pre-amplifier stages to the PGA issignificantly reduced by incorporating the use of a clock-lesspre-amplifier stage. Since the clocking occurs after the pre-amplifierstage in the capture and memory regeneration stages, the capture stageisolates the clock kick-back from proceeding into the pre-amplifierstage. In addition, because the pre-amplifier stages are not clocked,their transient response of being enabled and disabled is eliminatedthereby reducing this portion of the clock kick-back of thepre-amplifier stage. Thus, the clock kick-back is inventively reducedeven when 17 comparators are driven simultaneously by one differentialsignal source by addressing several the above aspects in this design.

The differential output of the PGA is loaded by the input differentialgate capacitance of 17 comparators. Differential metal layer traces areused to interconnect these 17 comparators to the PGA. Both the inputcapacitance of the comparators and the capacitance of the metal layertrace add together to increase the differential capacitive load. Thedifferential capacitive load presented to the PGA is quite large andreduces the bandwidth of this signal path. Thus, this implies that theinput differential pair of transistors in the comparator should not beexcessively large in width in order to minimize the input capacitanceand reduce the corresponding power dissipation. However, the embodimentpresented in this inventive idea allows the input gates of thecomparator to have large width transistors which overcomes theperformance degradation due to capacitance mentioned above in the ADCcomparator. The inventive idea uses an active negative-capacitor circuitto cancel the effect of the large input capacitance of the comparators.This cancelation minimizes the capacitance between the PGA and ADC andextends the gain characteristics of the interface between the PGA'soutput and the inputs of the first stage of the comparators. The activenegative-capacitance, basically, is a cross pair NMOS with a capacitorconnecting their sources, and each NMOS is biased by a current source.

Various embodiments and aspects of the inventions will be described withreference to details discussed below, and the accompanying drawings willillustrate the various embodiments. The following description isillustrative of the invention and is not to be construed as limiting theinvention. Numerous specific details are described to provide a thoroughunderstanding of various embodiments of the present invention. However,in certain instances, well-known or conventional details are notdescribed in order to provide a concise discussion of embodiments of thepresent inventions.

BRIEF DESCRIPTION OF THE DRAWINGS

Please note that the drawings shown in this specification may notnecessarily be drawn to scale and the relative dimensions of variouselements in the diagrams are depicted schematically. The inventionspresented here may be embodied in many different forms and should not beconstrued as limited to the embodiments set forth herein. Rather, theseembodiments are provided so that this disclosure will be thorough andcomplete, and will fully convey the scope of the invention to thoseskilled in the art. In other instances, well-known structures andfunctions have not been shown or described in detail to avoidunnecessarily obscuring the description of the embodiment of theinvention. Like numbers refer to like elements in the diagrams.

FIG. 1A illustrates the circuit schematic of a conventional comparator.

FIG. 1B presents a portion of the circuit schematic of a conventionalcomparator and the coupling capacitance between the terminals of thetransistors pertaining to clock kick-back.

FIG. 2 depicts the timing diagrams of a conventional comparator.

FIG. 3A shows the circuit schematic of the comparator and post operatingblock in accordance with the present invention.

FIG. 3B illustrates the symbol of FIG. 3A in accordance with the presentinvention.

FIG. 4 presents block diagram of the comparator in accordance with thepresent invention.

FIG. 5A illustrates the circuit schematic of the innovative comparatorin accordance with the present invention.

FIG. 5B depicts the reset pulse generator used in the circuit schematicof FIG. 5A in accordance with the present invention.

FIG. 5C shows a table illustrating the voltage inputs and outputs of theinventive circuit of FIG. 5A in accordance with the present invention.

FIG. 5D illustrates a portion of the circuit schematic of the innovativecomparator of FIG. 5A illustrating the two series coupling capacitancereducing the clock kick-back in accordance with the present invention.

FIG. 6A presents a simulation of the timing for a comparator circuit inabsence of transistors M₂₁, M₂₂, M₂₃ and pulse generator in accordancewith the present invention.

FIG. 6B illustrates a simulation of the timing for a comparator circuitof FIG. 5A in accordance with the present invention.

FIG. 7 depicts a diagram of the resistor ladder and comparators inaccordance with the present invention.

FIG. 8A shows a conventional transistor layout of the current source andone of the input pairs of two comparators.

FIG. 8B presents an inventive transistor layout of the current sourceand one of the input pairs of two comparators in accordance with thepresent invention.

FIG. 5C depicts an inventive transistor layout of the current source andboth input pairs of two comparators in accordance with the presentinvention.

FIG. 9 illustrates a diagram of the resistor ladder and comparators withactive negative-capacitance in accordance with the present invention.

FIG. 10A depicts the equivalent circuit diagram of the activenegative-capacitance in accordance with the present invention.

FIG. 10B illustrates the equivalent circuit diagram of the single-endedversion of the active negative-capacitance in accordance with thepresent invention.

FIG. 11 shows a Process, Voltage and Temperature (PVT) current sourceapplied to the active negative-capacitance of the in-phase I andquadrature phase Q channels in accordance with the present invention.

FIG. 12 presents a plot of the frequency response of the PGA and ADCwith and without the active negative-capacitance circuit in accordancewith the present invention.

DETAILED DESCRIPTION OF THE INVENTION

The inventions presented in this specification can be used in any wiredor wireless system or any low power supply voltage design. Thetechniques are applicable to any amplifier design, ADC design, or PGAand ADC interface design. These techniques can be extended to othercircuit designs where an increased bandwidth between two interfaces, aclock kick-back reduction, or a matched transistor within a circuit isrequired.

A comparator that is clocked in the first pre-amplifier stage isillustrated in FIG. 1A. The basic construction of the clockedpre-amplifier stage includes a ground switch M₁ with a gate coupled to aclock CK. The drain of M₁ 1-9 is coupled to the source of two N-channeltransistors M₂ and M₃. M₂ is driven by V_(IN−) while M₃ is driven by theother differential input signal V_(IN). The drain of M₂ is coupled to1-1 and is also coupled to the drain of P-channel transistor M₇controlled by the same clock CK. The drain of transistor M₃ 1-2 iscoupled to the drain of P-channel transistor M₁₂ controlled by the sameclock CK. A RAM cell is coupled between the two nodes 1-1 and 1-2 andthe supply VDD. The transistors of the RAM cell include M₄, M₅, M₉ andM₁₀. Note that M₄ is cross coupled to M₅ and M₉ is cross coupled to M₁₀.The drain of N-channel transistor M₄ is coupled to the drain ofP-channel transistor M₉. The drain of N-channel transistor M₅ is coupledto the drain of P-channel transistor M₁₀. The two outputs of the RAMcell 1-3 and 1-4 are also coupled to transistors M₈ and M₁₁ andcontrolled by the same clock signal CK. In addition, the two outputs ofthe RAM cell are coupled by the P-channel transistor M₆ controlled byclock CK to initialize the cell. Thus, this first stage of thepre-amplifier stage uses a single clock to initialize and capture thesignal being presented at the two input nodes V_(IN−) and V_(IN+).

Basically, with this topology, when the clock CK flips from low to high,the tail transistor (M₁) will drag the sources of the two inputtransistors to ground rapidly, leading to a large kick back to the inputsignal through C_(gs2) and C_(gs3) (see FIG. 2 b) of the transistors M₂and M₃, respectively, and disturbs the operation of other comparators.When the clock flips from high to low, the drain of the two inputtransistors will be pre-charged to VDD, also causing kick back at theinput signal through C_(gd2) and C_(gd3). These kick-backs become moreserious when the comparator is operated at higher frequencies (forexample, 2.64 GHz).

In the second portion of the circuit, the outputs of the clockcomparator 1-3 and 1-4 are applied to the inverters 1-5 and 1-6. Theseinverters drive the gates of the N-channel transistors M₁₃ and M₁₄,respectively. These two N-channel transistors rewrite or maintain thecontents of the data that is stored in the cross coupled memory cellcomposed of inverters 1-7 and 1-8. The outputs are drawn from the outputof this coupled cross coupled memory cell consisting of the two back toback inverters and these outputs are the V_(N1−) and the V_(P1∘).

Looking at the clocked pre-amplifier stage, a P-channel transistorcouples the outputs 1-3 and 1-4 together to initialize the cell when CKis low. This transistor is labeled as M₆ and is clocked by CK. When CKgoes low, the two outputs of the differential comparator equalizesimultaneously. When clock CK is low, the N-channel transistor M₁ isdisabled and all of the remaining P-channel transistors M₇, M₈, M₉, M₁₀,M₁₁ and M₁₂ are all enabled causing the nodes 1-3 and 1-4 to pre-chargeto VDD. Once the clock goes high enabling M₁, all of the P-channeltransistors M₆-M₁₂ become disabled and the contents of the first RAMmemory cell consisting of the cross coupled transistors M₄, M₅, M₉ andM₁₀ amplifies the difference of the signals that is applied to theN-channel gates M₂ and M₃. A transient behavior occurs before the cellcan make a decision. This transient behavior occurs because both outputnodes were pre-charged to VDD in the initialization state. When the cellbecomes enabled, a transient occurs until the circuit reaches asteady-state and finally captures the input signal, at this point, thevoltages at nodes 1-1, 1-2, 1-3 and 1-4 are stable. Once this steadystate occurs, the first RAM memory cell stabilizes the voltages at itsoutput nodes 1-3 and 1-4 and the captured information is then applied tothe second stage portion of the latch. The inputs applied to M₂ and M₃are amplified and captured in the RAM cell. As the differential voltagebetween these inputs decreases, the kick-back generated by thispre-amplifier and the others may produce erroneous results at nodes 1-3and 1-4 before the pre-amplifier has had a chance to stabilize. Thesecond portion comprises the two inverters 1-5 and 1-6 used to amplify,isolate and provide the signal to a second RAM memory cell consisting ofinverters 1-7 and 1-8 enabled by M₁₃ and M₁₄.

The buffer interface provides coupling of the differential signal fromthe first RAM cell to the second RAM memory cell. The second RAM memorycell comprises the back to back coupled inverters 1-7 and 1-8. Thecontent of the second RAM memory cell is either over-written with newcontent or maintains the same content depending on the current valuesbeing introduced to the cell and the previous values held by the cellfrom the last capture. The outputs of the comparator are the voltagesV_(N1−) and V_(P1∘). The transistors M₁₃ and M₁₄ of the second RAMmemory cell over-write the contents of the cell if the transistor thatis enable by one of the inverter's 1-5 or 1-6 output flip the contentsof the second RAM cell. Note that during the first RAM cellinitialization, nodes 1-3 and 1-4 are high preventing M₁₃ and M₁₄ fromoverwriting the second RAM cell. The transistors M₁₃ and M₁₄ of thesecond RAM memory cell maintain the contents of the cell if thetransistor that is enable by the inverter's output couples the logicalzero contents of the second RAM cell to ground. The transistors M₁₃ andM₁₄ of the second RAM memory cell switch the contents of the cell if thetransistor that is enable by the inverter's output couples the logicalone contents of the second RAM cell to ground.

The circuit in FIG. 1A suffers several drawbacks; 1) the ground switchtransistor M₁ is in series between VDD and the memory cell reducing thevoltage headroom and decreasing the performance; 2) the transientbehavior of the first RAM cell from initialization to steady stateintroduces large voltage swings at nodes 1-1, 1-2 and 1-9 causingpotential capacitance signal feedback between the terminals of thetransistors M₂ and M₃ into the input signal which introduces a kick-backinto the input signal thereby causing inaccuracies in the capture of thesignal; 3) the clocked transistors M₁ and M₆-M₁₂ introduce a kick-backinto the internal nodes of the comparator effecting the accuracy of thecapture of the input signal; and 4) the changing clock edge on the gatesof M₁, M₇ and M₁₂ introduce a clock kick-back into the input signal anddecrease the accuracy of the captured signal. Lastly, the circuitrequires a larger clock driver to drive the capacitive load of all thetransistors thereby increasing the overall power dissipation of thefinal circuit.

FIG. 1B illustrates a simplified version of the schematic in FIG. 1A andshows the various capacitors between the terminals of a transistor thatinfluence the internal nodes due to clock kick-back that affects thebehavior of the circuit. The clock CK swings nearly the nearly full rail(VDD to VSS) and any capacitance coupled to the clock line transferssome of that clock to the other side of the capacitor. The gate to draincapacitance C_(gd1) in transistor M₁ and the gate to source capacitancesC_(gs2) and C_(gs3) of transistors M₂ and M₃, respectively, areillustrated in FIG. 1B. In addition, the gate to drain capacitanceC_(gd2) and C_(gd3) of M₂ and M₃ and the gate to drain transistorcapacitance C_(gd7) and C_(gd12) of M₇ and M₁₂, respectively, are alsodepicted. These coupling capacitors help illustrate how the clockkick-back functions. When the differential comparator is enabled, theclock signal CK transitions from a zero to a one and injects a charge atnode 1-9 via C_(gd1). The injected charge at 1-9 is also coupled to theinput nodes V_(IN−) and the V_(IN∘) via C_(gs2) and C_(gs3),respectively. Fortunately, these two capacitors are in series somewhatdiminishing the clock kick-back. However the clock that is applied toP-channel transistors M₇ and M₁₂ is also fed through the gate to drainnode to nodes 1-1 and 1-2, respectively. The voltage at these two pointsis also coupled to the input via gate to drain capacitance of thetransistors M₂ and M₃ or C_(gd2) and C_(gd3). So thus, as the clockvaries from one polarity to another, the injected charge of the clocksignal is fed back to the input node and is a first portion of the clockkick-back.

A second portion of the clock kick-back is the transient behavior of thepre-amplifier stage from initialization to steady state which introduceslarge transient voltage swings at nodes 1-1, 1-2 and 1-9. The transientvoltage signal swing at node 1-1 is coupled to the input node V_(IN−) bya capacitor C_(gd2), the transient voltage signal swing at node 1-2 iscoupled to the input node V_(IN+) by a capacitor C_(gd3) and thetransient voltage signal swing at node 1-9 is coupled to the input nodeV_(IN+) and V_(IN−) by the capacitor C_(gs3) and C_(gs2), respectively.These feedback capacitance signals between the terminal of thetransistors M₂ and M₃ into the input nodes accounts for a second portionof the of the clock kick-back and introduces inaccuracies in the captureof the signal.

The advice results due to the clock kick-back of certain nodes for thecomparator depicted in FIG. 1A are illustrated in the FIG. 2. The topwaveform V_(ID) is an ideal plot of the input signal. The ideal signalillustrates what the input signal V_(IN+) would appear without a clockkick-back. The middle and bottom waveforms V_(IN+) and CK are thesimulated results of the input signal V_(IN+) and CK. This CK waveformpresents an edge 2-2 going high which enables the clocked comparator anddue to the effects that were mentioned earlier, the ideal input signalexperiences of glitch at 2-1 as shown in the actual input signalV_(IN+). Similarly, when the CK waveform goes low 2-4, the actual inputsignal V_(IN+) experiences a positive glitch 2-3 due to the effects thatwere mentioned earlier. These glitches occur each time the clock makes atransition because of the coupling capacitance due to the capacitancesbetween the terminals of the transistor and the transient behavior ofthe comparator being enabled and disabled. The V_(IN+) signal followsthe ideal input signal V_(ID) with the addition noise or glitches due tothe clock kick-back. The extraction of the data from the input signalV_(IN+) becomes more difficult due to these glitches and increases theinaccuracy of the translation of the analog to digital conversion. Thus,a source of error of determining the voltage values of the input signalbasically occurs due to the clock kick-back. The kick-back isapproximately 200 mV in the positive and the negative directions whichcauses errors to occur when one wants to capture the ideal input signal.If the kickback clock effect can be decreased or reduced then theaccuracy of the translation of the signal would improve.

To minimize the above issues, two major embodiments are introduced. Thefirst is to remove the clock transistors in the pre-amplifier stage, sothat the pre-amplifier will keep amplifying the signal without beingdisturbed by a clock kick-back signal and not suffer the large transientbehavior of being enabled and disabled. The second is to use a resetpulse generator circuit to create a “reset” signal applied to a resettransistor in the capture stage to initialize the regeneration stage.The generator also creates a “pass” signal applied to pass transistorsin the capture stage that allows the signal from the pre-amplifier topass to the regeneration stage with proper timing. The “reset” signaland “pass” signal are adjusted to minimize the kick-back.

A differential comparator which eliminates the clocking in thepre-amplifier stage of the comparator is illustrated in FIG. 3A. Thesecond stage latching with a clock CK occurs in the Post ClockingOperation block 3-5. Depending on when the second latch is opened incapturing the data will make an influence on how accurate the capturesignal corresponds to the actual or ideal signal. The circuitconfiguration of the pre-amplifier stage of the clock-less comparatorusing transistors and loads is illustrated. The output of this circuitcouples to the Post Clocking Operation block 3-5 via lines 3-3 and 3-4.A biasing voltage V_(B) is applied to transistors M₁₃ and M₁₆ whichmirror a current source. The drain of transistors M₁₅ 3-1 and M16 3-2are each connected to a differential transistor circuit. The firstdifferential transistor circuit connected to 3-1 consists of N-channelsM₁₇ and M₁₈ coupled through loads to VDD. The loads can be comprised ofresistors and/or reactive components. This first differential transistorcircuit has the input signals V_(IN−) and V_(RB) applied to the gates ofM₁₇ and M₁₈, respectively. The second differential transistor circuitconnected to 3-2 is also coupled through the same loads to VDD. Thesecond differential transistor circuit consists of the N-channels M₁₉and M₂₀ connected to node 3-2. This second differential transistorcircuit has the input signals V_(RA) and V_(IN+) applied to the gates ofM₁₉ and M₂₀, respectively. The voltages V_(RA) and V_(RB) are inputreference signals and are derived from a resistor chain (to be describedshortly). The right leg of both differential pairs sinks current fromload Z₂ while the left leg of the differential pair sinks current fromload Z₁ where both of the loads Z₁ and Z₂ are coupled to VDD. Thetransistor structure of M₁₆, M₁₉ and M₂₀ is called a differential stage3-7. This differential stage 3-7 has a current mirror, M₁₆, a firstinput transistor, M₁₉, and a second input transistor, M₂₀. The outputsof the differential stage are coupled to nodes 3-3 and 3-4. Thetransistors M₁₅, M₁₇ and M₁₈ make a second differential stage. The firstoutput 3-3 of the pre-amplifier stage is extracted at the drains oftransistor M₁₈ and transistor M₂₀ and the second output 3-4 of thepre-amplifier stage is tapped at the drains of transistor M₂₀ andtransistor M₁₇. These two outputs are differential outputs. Note thatthis pre-amplifier stage is not clocked at all, in other words, it isclock-less. This should reduce the clock kick-back and improve thecapture of the input signal.

All of the clocking for this innovative comparator is applied to thePost Clocking Operation block 3-5 to generate the outputs V_(P1+) andV_(N1−). The post clocking operation segregates the clocking from thepre-amplifier stage of the differential comparator and minimizes theclock kick-back. The symbol of the differential comparator isillustrated in FIG. 3B. It has four input signals and two outputsignals. The biasing voltage and power supply voltages are notillustrated in this symbol; however, the two input signals V_(IN+) andV_(IN−) are shown on the inside of the four of the inputs on the leftand the outer inputs are the two input reference voltages V_(RA) andV_(RB), the outputs of the differential comparator 3-6 is provided atthe nodes V_(P1+) an V_(N1−).

FIG. 4 illustrates a block representation of the differential comparator3-6 illustrated in FIG. 3B and FIG. 3A. The pre-amplifier stage 4-1 isdriven by the four input signals which are the two reference signalsV_(RA) and V_(RB) as well as the input signals V_(IN+) and V_(IN−). Theoutput of the pre-amplifier stage 4-1 corresponds to the two previousoutputs 3-3 and 3-4 illustrated in FIG. 3A. The remaining blocks are theblocks that are within the Post Clocking Operation block 3-5 shown inFIG. 3A. The clock is applied in the upper left to the Reset PulseGenerator 4-3 and generates two outputs that are applied to the resetand pass transistors of the Capture stage and Initialize Memory block4-2. This block captures the outputs of the pre-amplifier stage on lines3-3 and 3-4 and applies the captured signals to buffers 4-4 and 4-5. Theclock is also applied to Clocked Memory Cell block 4-6 which capturesthe outputs of the two buffer circuits. The Clocked Memory Cell block4-6 is then applied to the Memory Cell block 4-7 to generate two outputsV_(P1∘) and V_(N1−). The clock signal is also applied to the PostClocking Operation block of other comparators.

A more detailed analysis of FIG. 4 is illustrated in FIG. 5A. Thepre-amplifier stage, the capture stage and memory regeneration, the twobuffers, and the latching memory stage are identified along the top ofthe FIG. 5A. The loads Z₁ and Z₂ are replaced with resistors R₁ and R₂,respectively. The reset pulse generator 4-3 of FIG. 4 is shownschematically in FIG. 5B. This circuit generates the timing of thewaveforms for the capture stage 5-1. The clock signal enables the resetpulse generator to generate pulses. The clock is applied to the firstinverter 5-4 which is buffered by 5-5. The output of 5-5 is CK1 and isalso applied to other portions of the capture stage and memory circuit.The clock CK1 is applied to inverter 5-6 which is coupled to inverter5-7. The signals CK1 and the output of inverter 5-7 are applied to theNOR gate 5-10 which is buffered by the inverter 5-12 to generate thepass pulse V_(P). Simultaneously, the clock CK1 and the output ofinverter 5-8 driven by inverter 57 are applied to a second NOR gate 5-9that is buffered by 5-11 to generate the reset pulse, V_(R). These twosignals, the pass and reset pulses are applied to the reset and passtransistors of the capture stage 5-1 which consists of transistors M₂₁,M₂₂ and M₂₃. The reset pulse, V_(R), is applied to reset transistor M₂₁to reset the memory cell consisting of transistors M₂₄-M₂₈. The two passtransistors M₂₂ and M₂₃ are clocked by the pass pulse, V_(P), andtransfer the contents of the pre-amplifier stage and pass them into thefirst clocked memory cell. Note that V_(P) and V_(R) are bothlow-effective since M₂₁, M₂₂ and M₂₃ are P-channels.

The memory regeneration stage consists of a common clocked transistorM₂₄ coupled to a first inverter M₂₇ and M₂₅ and a second inverter M₂₈and M₂₆. The output of the first inverter is coupled to the input of thesecond inverter. Similarly, the output of the second inverter is coupledto the input of the first inverter forming a back-to-back coupled RAMcell that is clocked by CK1 via the transistor M₂₄. The two outputs ofthe RAM cell 5-2 and 5-3 are buffered by the buffers 1-5 and 1-6,respectively. The inverter's output is then applied to a latching memorystage consisting of the back-to-back inverters 1-7 and 1-8. Each outputof the latching memory stage has an N-channel transistor M₂₉ and M₃₀coupled to ground. Depending on the values of the differential signalprovided by the inverters 1-5 and 1-6, the contents of the latchingmemory stage can be switched or maintain at the same values as before.The outputs of the latching memory cell are V_(P1∘) and V_(N1−).

At beginning of each clock cycle (starting from the falling edge), asclock goes low, the memory regeneration stage is disabled. After a fixeddelay, a reset pulse is generated, initializing both outputs of thememory regeneration stage. The nodes 5-2 and 5-3 are equalized and areabove the threshold voltage of the inverters 1-5 and 1-6 preventing thelast RAM cell, 1-7 and 1-8, from being overwritten. This stepeffectively reduces the chance of error caused by the kick-back. Afterthe reset pulse is disabled, the pass pulse is generated, shorting thepre-amplifier output and memory regeneration stage. Depending on thepulse width of this pass pulse, a slight kick-back can still be observedat pre-amplifier output. Nevertheless, since the memory regenerationstage output has been initialized, the kick back will not cause thememory regeneration stage output to flip to the wrong polarity. Inpractice, however, multiple kick-backs from different comparators mayfeed through to the input node of the pre-amplifier stage and impair itscorrectness.

The dotted box 5-13 of FIG. 5C illustrates two conditions for thecomparator given in FIG. 5A. The first condition listed isV_(IN+)−V_(IN−) is greater than the difference of reference voltageV_(RA)−V_(RB), then the output of the circuit V_(P1+) would be a logicalone. The second condition listed is V_(IN+)−V_(IN−) is less thanV_(RA)−V_(RB), then the output of the circuit V_(P1+) would be a logicalzero.

The coupling capacitances between the terminals of the transistors in aportion of the circuit FIG. 5A are illustrated in FIG. 5D. The V_(P)pulse is generated by the clock signal CK1 and this signal swings railto rail and can cause clock kick-back through transistor M₂₂ via gate tosource capacitance C_(gs22) and the drain to gate capacitance of M₂₀.C_(gd20) transfers the signal on 3-3 and passes it to the input signalV_(IN+). Note that there are two series capacitors so that the signal ofthe clock V_(P) is diminished before it is sensed at the V_(IN+)terminal. To achieve a comparable performance as when the clock andchannel transistor were located at the base of the clock comparator asin FIG. 1A, the transistor widths in FIG. 5A can be reduced for thecircuit illustrated. This reduction in width minimizes the overlapcapacitance between the terminals of the transistors which is anotherreason for decreasing any type of clock kick-back that would be sensedat the input voltages of V_(IN+). Furthermore, the pre-amplifier stageis in the steady state condition eliminating a portion of the clockkick-back since this circuit did not need to be initialized. Thesimulation plots of the circuit illustrated in FIG. 5A are presented inFIG. 6A and FIG. 6B.

The minimization of the clock kick-back is illustrated between FIG. 6Aand FIG. 6B. In FIG. 6A, three waveforms are shown; the top is the idealsignal V_(ID), the middle signal is CK1 and the bottom signal isV_(C)=V_(E) because the capture stage 5-1 in FIG. 5A has been replacedby a short. Note that V_(C) experiences a clock kick-back of over 700millivolts due to the clock CK1 making a transition. This clockkick-back of the internal node could introduce a significant voltagevariation on the input node V_(IN+) which would affect the ideal signalbeing applied.

To reduce the kick-back, two major modifications are introduced: 1) Theclocked transistors in the pre-amplifier stage are removed, so that thepre-amplifier stage will keep amplifying the signal without beingdisturbed by clock kick-back; and 2) A reset pulse generation circuit(shown in FIG. 5B) is created to generate carefully positioned reset andpass pulses to control the innovative comparator.

In FIG. 6B, the capture stage 5-1 replaces the short mentioned for thesimulation results of FIG. 6A and the top waveform is CK1. The next twowaveforms correspond to the waveforms of the reset pulse V_(R) and thepass voltage pulse V_(P), respectively. The reset pulse generationcircuit generates the “reset” pulse for the reset transistor of thecapture stage and the “pass” pulse for the pass transistors of thecapture stage. The reset pulse initializes the regeneration stage whilethe pass pulse transfers the signal from the pre-amplifier stage to theregeneration stage with proper timing via the pass transistors in thecapture stage. The next waveform is the V_(C) signal which experiences avery small kickback of 100 millivolts or so while the bottom waveform isa difference waveform V_(DF). The difference waveform V_(DF) shows thedifference between the waveform V_(E) and the waveform on node 5-2. TheV_(C) V_(E) and 5-2 nodes are labeled in FIG. 5A. The lowest waveform isapplied to the inverter 1-6 that passes the signal to the latchingmemory stage which generates a rail-to-rail voltage. This waveform isapplied to the latching memory stage to generate the final output of thecomparator.

Note that the kick-back in FIG. 6B has been significantly decreased whencompared to the results of FIG. 6A. The rising edge of V_(R) is adjustedin comparison to the falling edge of V_(P) within a given time window tominimize the kick-back. This can be achieved by appropriately sizing theinverters/gates of the reset pulse generation circuit in FIG. 5B. Thistiming adjustment ensures a certain amount of hold time has beensatisfied for the first clocked memory cell consisting of transistorsM₂₄ through M₂₈. Such an implementation leads to the control signalwaveforms (V_(P) and V_(R)) shown in FIG. 6B. It can be seen that thememory regeneration stage is enabled (CK1 goes high) before the passpulse is disabled (V_(P) goes high). This is based on the observationthat the slow memory regeneration stage will not change thepre-amplifier output significantly. Since the pre-amplifier will haveanother half clock cycle to sample the input signal, this overlap is notlikely to cause error for the next sample. Also, FIG. 6A illustratesthat without isolation/reset pulse, the pre-amplifier stage output(V_(C)=V_(E)) would be totally distorted by the generated kick-back.

The 4-bit flash ADC in FIG. 7 consists of 17 comparators, 15 of whichdivide a reference voltage into 16 sections, while the other twoindicate overflow/underflow. To generate 4-bit binary code, the analoginput is divided into 2⁴=16 levels, which requires only 15 comparators(comparator #2˜#16), so the outputs of comparator #1 and #17 are justoverflow/underflow indicators. The thermometer code generated by thecomparator array passes through a bubble cancellation circuit and isthen translated into binary code, which is de-serialized by thefollowing stage from 2640 MSa/s to 220 MSa/s (Mega Samples per sec).

The core concept of this ADC is the high-speed fully-differentialcomparators which are clocked at 2640 MHz. Basically, each comparatorconsists of four parts: a pre-amplifier stage which samples andamplifies the input signal from preceding stage (PGA); a capture stage;a regeneration stage with cross-coupled pairs that is clocked toregenerate the small signal and amplifies the signal to the next stage;and a latching stage which latches up the comparison results after beingregenerated providing the signal to the following digital CMOScircuitry.

To work at a 2640 MHz clock rate, the comparators must provide high dcgain to regenerate the signal within the allowed time period yetminimize metastability issues. Fast regeneration, on the other hand,leads to strong kick-back noise at the input node of the pre-amplifier,and due to the Miller feedback effect the noise potentially results infalse decisions when the input signal applied to the pre-amplifier stageis small. In addition, large input transistors are also susceptible toclock kick-back when the pre-amplifier stage is clocked as in usualimplementations.

A comparator with large width transistors can operate quickly but sincethe PGA can be loaded with 17 comparators directly, the inputcapacitance of the comparators can be quite large thereby slowing downthe output of the PGA. In addition, the power constraint would beexceeded if large width transistors were used in the pre-amplifierstage.

The innovative comparator circuit illustrated in FIG. 5A with reducedclock kick-back is utilized 17 times in the ADC which is illustrated inFIG. 7. This ADC is a flash converter because all 17 comparators operatesimultaneously to calculate the translation. Since all comparatorsoperate simultaneously, the clock kick-back into the input signalsV_(IN+) and V_(IN−) is increased 17 times that of a single comparator.This potentially introduces a multiplicative effect at the input signalsdemonstrating the importance of reducing the clock kick-back to thelowest possible level in the comparator since this comparator is usedmultiple times.

Instead of having a resistor ladder which generates 33 referencevoltages, a resistor ladder having 16 resistor segments is used, and theconnections between the resistor segments to the input of thecomparators are unconventional. Basically, the implementation issymmetrical with respect to V_(R8). V_(IN+) can be either higher orlower than V_(IN−), and the point where V_(IN+)=V_(IN) is set at theboundary of output V_(P8).

The two inner input signals V_(IN+) and V_(IN−) of the comparator inFIG. 7 are connected to the outputs of the programmable gain amplifierfrom the proceeding circuit. The outer input reference signals aretapped into the resistor chain formed of 16 resistors 7-2 through 7-9.Note the . . . on all of these lines indicating that there are moreresistors and comparators within that region. The resistor chain ispositioned between VDD and a current source I₁ which is connected toVSS. This resistor string provides a segmented voltage division betweenVDD and V_(th)+Δ which can be adjusted by the current I₁.

The clock generation circuit works identically in each comparator. Thelocal clock generation circuits avoid extra clock jitter from beinggenerated. Decreasing the jitter improves the performance of the ADC.

As described earlier, there are 17 comparators and the first and lastcomparators coupled to the resistor string are used for underflow andoverflow evaluation. The negative outer input of the underflowcomparator (Comp #0) is connected to VDD while the positive outer inputis connected to V_(th)+Δ. Thus, the resistor chain provides two inputreference signals or voltages to each comparator. The negative outerinput of the overflow comparator (Comp #16) is connected to V_(th)+Δwhile the positive outer input is connected to VDD. When the inputsignals, V_(IN+) and V_(IN−), remain within the range between V_(th)+Δand VDD, comp #0 is at a logical high (1) and com #16 is at a logicallow (0) indicating no underflow or no overflow, respectively. However,when the difference between input signals, V_(IN−) and V_(IN+), isgreater than the bound of (VDD−V_(th)−Δ), comp #16 is set to a logicalhigh (1) which indicates an overflow. And, when the difference betweeninput signals, V_(IN+) and V_(IN−), is less than the bound of−(VDD+V_(th)+Δ), comp #0 is set to a logical zero (0) which indicates anunderflow.

The remaining comparators (#1-#415) are used to digitize the analogsignal which remains within the bounds of V_(th)+Δ and VDD. For example,comparator #1's outer negative terminal is connected to the top ofresistor 7-3 which is the voltage V_(R15) and its outer positiveterminal is connected to V_(R0) at the lower end of resistor 7-9 in theresistor string. This comparator generates V_(P1). Similarly, comparator#15 which generates V_(P15) has its outer positive terminal connected toVDD at the top of resistor V_(R16) and its outer negative terminalconnected to the bottom of V_(R0) in the resistor string. The outputs ofthese comparators starting from Comp #15 to Comp #1 would then generatea 1 followed by a number of zeros and the division between one and zerois dependent on the input voltage of V_(IN+) and V_(IN−). For example,in dotted box 7-1 if V_(IN+)−V_(IN−) is greater than V_(R15)−V_(R1) thenthe output of comparator #15 V_(P15+) is equal to a digital one. On theother hand, if V_(IN+)−V_(IN−) is less than V_(R15)−V_(R1) then theoutput at comparator #15 V_(P15+) is equal to a digital zero. As theinput voltage increases, more ones are added to the digital string. Thebubble cancellation translates the string into a 4-bit digital binarysignal.

As the clock signal propagates through all comparators, “bubbles” mayappear at the output thermometer code due to different clock delays. Abasic bubble cancellation circuit following the comparator array cancompensate for this effect. Basically, for each thermometer code, ittakes 3 different thermometer code outputs that correspond to 3consecutive levels. If the two higher levels are both a “0” and a “I”corresponds to the lowest level, then a new thermometer code “1” isgenerated corresponding to the lowest level only if the higher twolevels are both “0”. For example, Vn10, Vn9, and Vp8 will go to the sameAND gate that generates a new thermometer code. In that case, when thereis a bubble at Vp9, meaning Vp10=1 (Vn10=0), Vp9=0 (Vn9=1), and Vp8=1(Vn8=0), the “1” at Vp8 will be discarded in order to remove the bubbleat Vp9.

The analog comparator contains differential circuitry which needs tocompare two different voltages. The closer these two voltages approachone another, the need of the differential circuit in the comparators todistinguish the small difference increases. Any non-uniformity in thedifferential circuit becomes more exposed during this criticaldistinction of the small voltage difference. A critical feature ofmaintaining uniformity is the matching of the transistors used in thedifferential circuit of the comparators. Transistor matching is aconcern during the fabrication of the transistors since localtopographical differences in the nearby environment of the transistorcan affect the forming of the transistor. Ideally, the local topographyshould be the same for each transistor and one way of achieving this isto place dummy transistors besides active transistors so that the localenvironment appears to be the same for the active transistor. However,the dummy transistors use up area on the die and increase the size ofthe circuit thereby increasing the cost and because of the greaterdistances decreasing the performance. In place of the dummy transistors,the innovative step is to abut the differential transistors togethersuch that the active transistor of one differential pair behaves as adummy transistor for a second differential pair.

The following issues and trade-offs emerged during the design process:Mismatches between transistors,

${{\Delta\; V_{TH}} = \frac{A_{VTH}}{\sqrt{WL}}},( {A_{VTH} = {4\text{∼}5\mspace{14mu}{mV}\text{/}{um}}} )$especially input transistor pairs, will lead to false output of thecomparator. The transistor has a width of W and a length of L. To keepthe mismatch well below 0.2 LSB (˜8 mV), with 60 nm channel length, awidth greater than 8 um is necessary.

The matching of transistors is better understood by the illustrations inFIG. 8A, FIG. 8B and FIG. 5C. FIGS. 8A and 8B show the layout of theinput transistors and current sources of two comparators. FIG. 5Apresents the layout using the conventional approach while FIG. 5B andFIG. 8C illustrate the layout of the embodiment with the inventivetechnique where the need for dummy transistors has been eliminated. Thisallows the transistors in both comparators of FIG. 8B and FIG. 8C to betightly packed without gaps providing a uniform environment for localprocessing. In this way, not only are the mismatches between differentcomparators minimized, but the layout also becomes more compact,allowing shorter routing distances for both the signal and the clock.

FIG. 8A illustrates the layout of a first portion of the differentialstage of the Nth comparator consisting of the current mirror driven byV_(B) and a differential pair driven by V_(IN+) and V_(RA). The layoutof a second portion of the differential stage of the [N+1]th comparatorconsisting of the current mirror driven by V_(B) and a differential pairdriven by V_(IN+) and V_(RA′). In order to ensure that all of thetransistors operate the same, due to local environmental conditions onthe integrated circuit, the dummy gates are inserted next to the activecircuitry to topologically alter the surface and structure such that alltransistors within the active circuitry area experience similar adjacenteffects. However, these dummy gates use up valuable semiconductor areaand cause the Nth comparator to be displaced farther from the [N+1]thcomparator.

An inventive improvement is to remove the intervening dummy gatesaltogether and place each comparator next to one another such that theactive transistor of the first differential stage becomes the dummytransistor for the second differential stage and vice versa. This isillustrated in FIG. 8B where now the first portion of the differentialstage of the Nth comparator abuts the portion of the differential stageof the [N+1]th comparator. Now the active transistor of the Nthcomparator is adjacent to the active transistor of the [N+l]thcomparator removing the requirement for dummy transistors. Thiseliminates the waste in area, decreases any wiring channels for clocksand other signals, and improves the performance of the circuit.

The complete transistor circuit for the Nth and [N+1]th comparators isillustrated in FIG. 8C. The lower section 8-1 is identical to the layoutillustrated in FIG. 8B. The upper section 8-2 illustrates the otherdifferential stages in the pre-amplifier stage of the comparator. Thedrains of the corresponding transistors in the upper and lower sectionsare connected together using metal 3 (M₃) and is not shown. The metal 3connection is well understood in the art and needs no furtherexplanation. The left half presents the transistor layout of the Nthcomparator including the input signals V_(B), V_(IN+), V_(IN−), V_(RA)and V_(RB). The right half presents the transistor layout of the [N+1]thcomparator including the input signals V_(B), V_(IN+), V_(IN−), V_(RA′)and V_(RB′).

To reduce the mismatches within one comparator and between comparators,all input transistors and their currents sources are put right next toeach other to serve as dummies of each other.

A folded resistor ladder is implemented to simplify routings fromresistor ladder to the differential comparators, with the price beingcomplicated routings to the bubble cancellation circuits. Comparatorssit next to each other to share transistor dummy fingers.

FIG. 9 illustrates an innovative circuit to improve the signal bandwidthtransfer between the proceeding programmable gain amplifier (PGA) to alldifferential inputs of the 17 comparators. The larger input gatetransistor area helps to minimize the mismatch condition of the ADCcomparator but introduces a larger input capacitance. An activenegative-capacitor circuit is used to cancel the effect of the largeinput capacitance of the comparators. This active negative-capacitancebasically is a cross-coupled N-channel pair with a capacitor couplingtheir sources together. The current in each N-channel transistor iscarried by one current source. Each of the comparators present an inputcapacitance at its V_(IN+) and V_(IN−) nodes. In addition, eachcomparator occupies an area on the semiconductor die and this area usageis multiplied 17 times. Thus, a large overall area is involved and inorder to propagate the output of the PGA signals to the input signalsV_(IN+) and V_(IN−) of all 17 comparators, the input signal requires atrace or metal interconnect between the PGA and each comparator. Thetrace introduces a significant amount of capacitance and thiscapacitance adds to the input capacitance of the comparators. Thedifferential interconnect includes a differential capacitive loadcomprising the capacitance of the interconnect, input capacitance of thecomparators and the drain capacitances of the PGA. The overallcapacitance causes the signal bandwidth to decrease in the interfacecircuitry between the PGA and the first stage of the comparator. This isa critical limiting feature for the performance of the system.

To overcome this short coming, the inventive cross couplednegative-capacitance circuit 9-1 of M₃₁ and M₃₂ illustrated in FIG. 9has been developed. The drain terminals of M₃₁ and M₃₂ are coupled tothe nodes V_(IN+) and V_(IN−). The cross coupling circuit senses thetransition of V_(IN+) and V_(IN−) and helps to speed up their transitionor shorten the time period of the transfer. The performance is furtherimproved by incorporating the two current sources 9-2 and 9-3 coupled toa supply, in this case a ground supply voltage or VSS. The capacitor C₁₀helps to stabilize the voltages at the sources of the two transistorsM₃₁ and M₃₂. Thereby, this circuit helps to speed up the transition andincreases the bandwidth at this critical interface juncture between theoutput of the PGA and the inputs to the 17 comparators.

An equivalent circuit representation of the cross couple circuit isillustrated in FIG. 10A. The input to the circuit is the voltage source10-1 which applies a current I_(IN) to the left portion of the circuitrepresenting the transistor 10-2 which has a g_(m)(V_(Y)−V_(S1)) andacross this current source is an impedance of r₀₁. The lower portion ofthe current source 10-2 is connected to R_(S1) at node V_(S1). The nodeV_(S1) is coupled to node V_(S2) via the capacitor C₁₀. On theright-hand side is the equivalent transistor of M₃₂ consisting of thecurrent source 10-3 with a g_(m)(V_(X)−V_(S2)) and a resistor r₀₂ inparallel. The top node V_(Y) is connected to the negative terminal ofthe input voltage 10-1. The transconductance operates on difference ofthe opposite voltage in the opposing leg with regard to the voltagewithin its own leg. As this voltage difference increases, thetransconductance increases helping to diminish the voltage differencethereby improving the bandwidth gain of this interface node. FIG. 10Billustrates the same circuit except the circuit is now single endedrepresentation where now the current source 10-5 represents thetransconductance gm(−V_(X)−V_(S1)) and is in series with the voltagesource 10-4 having a voltage of V_(IN)/2. The lower end of the currentsource 10-5 is coupled to ground via R_(S) and capacitor 2C₁₀.

By solving the small signal equivalent circuit in FIG. 10A, theassumption is that r_(o1)=r_(o2)=r₀, R_(S1)=R_(S2)=R₈, and C₁₀=C. Theequivalent impedance looking into this circuit is:

${\frac{Vin}{Iin}(s)} = {- \frac{2( {{ro} + {Rs} + {{gm}*{ro}*{Rs}} + {2\;{ro}*{Rs}*{sC}}} )}{( {{{gm}*{ro}} - 1} )( {{2\;{Rs}*{sC}} + 1} )}}$Neglecting current source impedance, we have:

${\frac{Vin}{Iin}(s)} = {- \frac{{{gm}*{ro}} + {2\;{ro}*{sC}} + 1}{{sC}( {{{gm}*{ro}} - 1} )}}$If we further neglect the channel length modulation of inputtransistors, it becomes:

${\frac{Vin}{Iin}(s)} = {{- \frac{2}{gm}} - \frac{1}{sC}}$The last equation illustrates that the impedance is dependent on boththe value of g_(m) and C.

The cross couple negative-capacitance circuit is used twice within thechip as depicted in FIG. 11. The first negative-capacitance circuit isapplied to the input voltages corresponding to the in-phase analogsignals of V_(IN+) _(—) _(I) and V_(IN−) _(—) _(I) while the secondnegative-capacitance circuit operates on the quadrature-phase analogsignals of V_(IQ+) _(—) _(Q) and V_(IQ−) _(—) _(Q). The two sets ofinput signals are provided by the differential outputs of two PGAs. Thecurrent source of the transistors is illustrated by transistor M₃₃ whichis coupled to a process, voltage and temperature digitally controlledanalog circuit which generates a current I_(R1). This circuit can becompletely analog controlled or completely digitally controlled but inthis case it uses a combination of the two controls to achieve thedesired value of I_(R1). The current mirror M₃₃ applies the voltage tothe gates of M₃₄, M₃₅, M₃₆ and M₃₇ providing a carefully controlledcurrent being applied to the drains 11-1, 11-2, 11-3 and 11-4 of thesetransistors. M₃₈, M₃₉ and C₁₃ and M₄₀, M₄₁ and C₁₄ are coupled to thesenodes as shown in FIG. 11. These two cross coupled negative-capacitancecircuits improve the performance between the programmable gainamplifiers (PGA) and the inputs of their corresponding ADC's. Acapacitor C₁₃ is placed between the sources of the first cross coupledcircuit formed by M₃₈ and M₃₉ while capacitor C₁₄ is placed between thesources of the second cross coupled circuit formed by M₄₀ and M₄₁.Although not shown, each of the individual cross couple circuits can becontrolled by separate and distinct analog control or current mirrors toperform additional functions if so desired.

In FIG. 12, the response at the output of the PGA driving 17 comparatorswithout the use of the innovative cross couple transistor is illustratedby the curve 12-1 and is measured with the squares. This curve has a cutoff frequency of about 0.88 GHz. When the innovative cross couplecircuit is used, the curve 12-2 illustrates the response of the circuitbetween the output of the programmable gain amplifier and the 17comparators. The curve shows a peaking of the response which pushes outthe bandwidth of the circuit 720 MHz to about 1.6 GHz. The gain at the1.5 DB points between both curves is depicted. This provides an improvedperformance at this critical interface. Thus, the signal bandwidthbetween the PGA and the ADC has improved by 720 MHz.

In this design, although the ADC itself has a 1 dB bandwidthapproximately 1.3 GHz (post-layout simulation), the bandwidth of the PGAdrops dramatically (2.6 dB drop at 880 MHz) when driving 17 comparatorsdirectly. There is a negative-capacitive component in this equivalentimpedance, which can be used to cancel the effect of the inputcapacitance of ADC and increase the bandwidth. FIG. 12 shows the effectof the negative-capacitance circuit in simulation. The 1.5 dB bandwidthof PGA increases from less than 880 MHz to 1.6 GHz with the help of thenegative-capacitance circuit.

A summary of some of the inventive apparatus for a clock-lesspre-amplifier system is provided.

A comparator apparatus comprising a first clock-less pre-amplifierstage, a capture stage coupled to the first clock-less pre-amplifierstage and a memory regeneration stage coupled to the capture stage,whereby the capture stage receives a reset and pass signals to transferdata from the first clock-less pre-amplifier stage to the memoryregeneration stage. At least one buffer is coupled to the memoryregeneration stage and a latching memory stage is coupled to the buffer.A reset pulse generator creates the reset and pass signals. A clockenables the memory regeneration stage and the clock also enables thereset pulse generator. A first differential stage of a first clock-lesspre-amplifier stage is abutted to a second differential stage of asecond clock-less pre-amplifier stage such that an active transistor ofthe first differential stage behaves as a dummy transistor for an activetransistor of the second differential stage. The first clock-lesspre-amplifier comprises: a first load coupled to a first output of afirst and a second differential stage, a second load coupled to a secondoutput of the first and the second differential stage, a first inputsignal and a first input reference signal coupled to the firstdifferential stage and a second input signal and a second inputreference signal coupled to the second differential stage where load canbe a resistive load.

An apparatus comprising a first load coupled to a first output of afirst and a second differential stage, a second load coupled to a secondoutput of the first and the second differential stage, a first inputsignal and a first input reference signal coupled to the firstdifferential stage, a second input signal and a second input referencesignal coupled to the second differential stage, the first outputcoupled to a third output by a first pass transistor, the second outputcoupled to a fourth output by a second pass transistor and the thirdoutput coupled to the fourth output by a reset transistor. The third andthe fourth output are coupled to a memory regeneration stage and thememory regeneration stage is coupled to at least one buffer. A thirddifferential stage is abutted to the second differential stage such thatan active transistor of the third differential stage behaves as a dummytransistor for an active transistor in the second differential stage. Alatching memory stage is coupled to the buffer. The first and secondpass transistors receive a pass signal to transfer data from the firstand the second output to the memory regeneration stage. The resettransistor receives a reset signal to initialize the third and thefourth output coupled to the memory regeneration stage.

A method of minimizing clock kick-back comprising the steps of couplinga first output of a first clock-less pre-amplifier stage to a first passtransistor, coupling a second output of the first clock-lesspre-amplifier stage to a second pass transistor, coupling the first passtransistor to a first input of a memory regeneration stage, coupling thesecond pass transistor to a second input of the memory regenerationstage, coupling a reset transistor between the first and second inputsof the memory regeneration stage, enabling the first and second passtransistor within a time window and adjusting the reset transistorwithin the time window to reduce the clock kick-back, thereby minimizingthe clock kick-back. The memory regeneration stage is coupled to atleast one buffer. The method includes abutting a second clock-lesspre-amplifier stage to the first clock-less pre-amplifier stage suchthat an active transistor of a first differential stage in the firstclock-less pre-amplifier stage behaves as a dummy transistor for anactive transistor of a first differential stage in the second clock-lesspre-amplifier and coupling a latching memory stage to the buffer. Thefirst and second pass transistors receive a pass signal to transfer datafrom the first output and the second output to the memory regenerationstage. The reset transistor receives a reset signal to initialize thefirst and the second output of the memory regeneration stage.

A summary of some of the inventive apparatus for a negative-capacitancesystem is provided.

A negative-capacitance apparatus comprising a first node coupled to adrain of a first transistor and a gate of a second transistor, a secondnode coupled to a drain of the second transistor and a gate of the firsttransistor, a capacitor coupled between a source of the first transistorand a source of the second transistor, a first current mirror coupledbetween a supply voltage and the source of the first transistor and asecond current mirror coupled between the supply voltage and the sourceof the second transistor. The apparatus also includes a first amplifierthat generates a differential signal coupled to the first and secondnodes. The first amplifier can be a programmable gain amplifier. Theapparatus also comprises a plurality of amplifiers that are driven bythe differential signal coupled to the first and second node. Each ofthe plurality of amplifiers comprises a pre-amplifier of a comparator. Acoupling is formed between the first amplifier and the plurality ofamplifiers. The pre-amplifier of the comparator is a clock-lesspre-amplifier. The pre-amplifier stages are abutted to one another suchthat an active transistor of a first differential stage in a firstpre-amplifier stage behaves as a dummy transistor for an adjacentdifferential stage in a second pre-amplifier stage.

A method of increasing a transfer bandwidth of a differential signalcomprising the steps of amplifying a differential input signal toprovide the differential signal driving a differential capacitive loadbetween a first and a second node, coupling the first node to a drain ofa first transistor and a gate of a second transistor, coupling thesecond node to a drain of the second transistor and a gate of the firsttransistor, coupling a capacitor between a source of the firsttransistor and a source of the second transistor, coupling a firstcurrent mirror between a supply voltage and the source of the firsttransistor, coupling a second current mirror between the supply voltageand the source of the second transistor and causing the differentialcapacitive load to be driven in a shorter time period, therebyincreasing the transfer bandwidth of the differential signal. A firstamplifier generates the differential input signal and a plurality ofamplifiers receives the differential input signal. The differentialcapacitive load comprises a differential capacitance of a differentialinterconnect, a differential input capacitance of the plurality ofamplifiers and a differential drain capacitance of the first amplifier.The first amplifier is a programmable gain amplifier. Each of theplurality of amplifiers is clock-less pre-amplifier of a comparator. Themethod includes abutting a plurality of clock-less pre-amplifier stagesto one another such that an active transistor of a first differentialstage in a first clock-less pre-amplifier stage behaves as a dummytransistor for an adjacent differential stage in a second clock-lesspre-amplifier stage.

An apparatus comprising a first amplifier coupled to a first and asecond node, a differential capacitive load coupled to the first and thesecond node, the differential capacitive load coupled between drains oftransistors in a cross coupled transistor circuit, a current sourcecoupled to a source of each transistor and a capacitor coupled betweenthe sources of the transistors. The apparatus also includes a pluralityof amplifiers coupled to the first and the second node and adifferential signal of the first amplifier drives the first and thesecond node. Each of the plurality of amplifiers is a clock-lesspre-amplifier of a comparator. The first amplifier is a programmablegain amplifier. The pre-amplifier stages are abutted to one another suchthat an active transistor of a first differential stage in a firstpre-amplifier stage behaves as a dummy transistor for an adjacentdifferential stage in a second pre-amplifier stage.

Finally, it is understood that the above descriptions are onlyillustrative of the principle of the current invention. Variousalterations, improvements, and modifications will occur and are intendedto be suggested hereby, and are within the spirit and scope of theinvention. This invention may, however, be embodied in many differentforms and should not be construed as limited to the embodiments setforth herein. Rather, these embodiments are provided so that thedisclosure will be thorough and complete, and will fully convey thescope of the invention to those skilled in the arts. It is understoodthat the various embodiments of the invention, although different, arenot mutually exclusive. In accordance with these principles, thoseskilled in the art may devise numerous modifications without departingfrom the spirit and scope of the invention. For example, the circuitshave a Doctrine of Equivalents, that is, P-channels transformed intoN-channels, VDD interchanges with VSS, voltages measured with respect tothe other power supply, the position of current sources moved to theother power supply, etc. The semiconductor die can include silicon,germanium, SI graphite, GaAs, SIO, etc. Although the circuits weredescribed using CMOS, the same circuit techniques can be applied todepletion mode transistors and BJT or bipolar circuits, since thistechnology allows the formation of current sources and source followers.When a transistor is specified, the transistor can be a transistor suchas an N-MOS or P-MOS. The CMOS or SOI (Silicon on Insulator) technologyprovides two enhancement mode channel types: N-MOS (N-channel) and P-MOS(P-channel) transistors or transistors. In addition, a network and aportable system can exchange information wirelessly by usingcommunication techniques such as Time Division Multiple Access (TDMA),Frequency Division Multiple Access (FDMA), Code Division Multiple Access(CDMA), Orthogonal Frequency Division Multiplexing (OFDM), Ultra WideBand (UWB), Wi-Fi, WiGig, Bluetooth, etc. The network can comprise thephone network, IP (Internet protocol) network, Local Area Network (LAN),ad hoc networks, local routers and even other portable systems.

What is claimed is:
 1. A method of improving a bandwidth response of aprogrammable gain amplifier (PGA) loaded by a plurality of analog todigital convertors (ADC's) comprising the steps of: coupling adifferential output of said PGA to a differential interconnect; couplinga differential input of each said plurality of ADC's to saiddifferential interconnect; and coupling a negative-capacitance circuitto said differential interconnect, wherein said negative-capacitancecircuit comprises: a first transistor cross-coupled to a secondtransistor with a capacitor coupling their sources together; a firstcurrent source coupled to said first transistor; and a second currentsource coupled to said second transistor.
 2. The method of claim 1,wherein said first current source and said second current source areboth controlled by one control circuit.
 3. The method of claim 1,wherein said first current source and said second current source havesubstantially equivalent characteristics.
 4. The method of claim 1,wherein said first current source and said second current source areeach controlled by separate control circuits.
 5. The method of claim 1,further comprising the steps of: adjusting a current of at least one ofsaid first current source and said second current source to improve saidbandwidth response of said PGA loaded by said plurality of ADC's.
 6. Themethod of claim 5, wherein said current of said first and said secondcurrent source is controlled by a digital adjustment or an analogadjustment in a control circuit.
 7. The method of claim 1, wherein saiddifferential output of said PGA is either a differential in-phase analogsignal or a differential quadrature-phase analog signal.
 8. An apparatuscomprising: a differential amplifier configured to provide adifferential analog signal on a first node and a second node; a firsttransistor coupling said first node to a first current node; a secondtransistor coupling said second node to a second current node, whereinsaid first transistor is cross coupled to said second transistor; athird transistor coupling said first current node to a supply voltage; afourth transistor coupling said second current node to said supplyvoltage; a capacitor coupling said first current node to said secondcurrent node; and gates of said third and fourth transistors coupled toa control circuit, wherein said output of said control circuit isconfigured to control a current in each of said third and said fourthtransistors.
 9. The apparatus of claim 8, further comprising: aplurality of comparators coupled to said first and second node.
 10. Theapparatus of claim 8, wherein said differential analog signal is eithera differential in-phase analog signal or a differential quadrature-phaseanalog signal.
 11. The apparatus of claim 8, further comprising: a fifthtransistor coupling said control circuit to said gates of said third andfourth transistors, wherein said control circuit is analog controlled,digitally controlled, or a combination of the two.
 12. The apparatus ofclaim 8, wherein said third transistor is matched to said fourthtransistor.
 13. The apparatus of claim 12, wherein said current in saidthird transistor substantially equals said current in said fourthtransistor.
 14. The apparatus of claim 8, wherein said differentialamplifier is a programmable gain amplifier.
 15. A method of improving abandwidth response of a programmable gain amplifier (PGA) loaded by aplurality of analog to digital convertors (ADC's) comprising the stepsof: coupling a differential output of said PGA to a differentialinterconnect; coupling a differential input of each said plurality ofADC's to said differential interconnect; coupling a negative-capacitancecircuit to said differential interconnect; sensing a transition of adifferential analog signal from said differential output of said PGA bysaid negative-capacitance circuit; and shorting a time period to performa transfer of said differential analog signal on said differentialinterconnect due to a response of said negative-capacitance circuit. 16.The method of claim 15, wherein said negative-capacitance circuitcomprises: a first transistor cross-coupled to a second transistor witha capacitor coupling their sources together; a first current sourcecoupled to said first transistor; and a second current source coupled tosaid second transistor.
 17. The method of claim 16, wherein said firstcurrent source and said second current source are both controlled by onecontrol circuit.
 18. The method of claim 16, wherein said first currentsource and said second current source have substantially equivalentcharacteristics.
 19. The method of claim 16, further comprising thesteps of: adjusting a current of said first and said second currentsource to further improve said bandwidth response of said PGA.
 20. Themethod of claim 19, wherein said current of said first and said secondcurrent source is controlled by a digital adjustment or an analogadjustment of a control circuit.